1. Field of the Invention
The present invention relates to the field of digital signal processing.
2. Description of the Background Art
So-called “Wireless Personal Area Networks” (WPANs) can be used for the wireless transmission of information over relatively short distances (between 1 cm, 10 m, and 120 m). In contrast to “Wireless Local Area Networks” (WLANs), WPANs require little or even no infrastructure for data transmission, so that small, simple, energy-efficient, and cost-effective devices can be implemented for a broad range of applications.
Standard IEEE 802.15.4 specifies, for example, low-rate WPANs, which are suitable with raw data rates up to a maximum of 250 kbit/s and stationary or mobile devices for applications in industrial monitoring and control, in sensor networks, in automation, and in the field of computer peripherals and for interactive games. In addition to a very simple and cost-effective implementability of devices, an extremely low power demand of devices is of critical importance for such applications. Thus, an objective of this standard is a battery life of several months to several years.
At the level of the physical layer, in the virtually globally available 2.4 GHz ISM band (industrial, scientific, medical) for raw data rates of fB=250 kbit/s, the IEEE standard 802.15.4 specifies a band spread (spreading) with a chip rate of fC=2 Mchip/s and an offset QPSK modulation (quaternary phase shift keying) at a symbol rate of fS=62.5 ksymbol/s.
In an 802.15.4 transmitter for the ISM band, the data stream to be transmitted is first converted to a sequence of PN sequences (pseudo noise) with the use of four databits in each symbol period (TS=1/fS=16 μs), in order to select a total of 16 PN sequences. Each symbol of four databits is assigned in this manner a symbol value-specific PN sequence of 32 PN chips (chip period TC=TS/32=500 ns=1 fC), which is transmitted instead of the four databits. The “quasi-orthogonal” PN sequences P0, P1, . . . , P15, specified in the standard, differ from one another in cyclic shifts and/or inversion of each second chip value (see IEEE Standard 802.15.4-2003, Chapter 6.5.2.3).
The PN sequences assigned to the successive symbols are joined together and then offset QPSK modulated (quaternary phase shift keying) by modulating, with half-sine pulse shaping, the even-indexed PN chips (0, 2, 4, . . . ) onto the in-phase (I) carrier and the odd-indexed PN chips (1, 3, 5, . . . ) onto the quadrature-phase (Q) carrier. To form an offset, the quadrature-phase chips are delayed by half a chip period TC with respect to the in-phase chips (see IEEE Std 802.15.4-2003, Chapter 6.5.2.4).
Both coherent and incoherent approaches are known to detect data symbols present in a receive signal. Whereas in coherent approaches the receive signal is converted into the complex envelope (baseband) with use of a carrier wave of the same frequency and phase and obtained from the carrier control circuit, in incoherent approaches at least the phase accuracy, within limits possibly also the frequency accuracy of the carrier wave, can be omitted.
A coherent receiving unit is known from the textbook “Nachrichtenübertragung” [Message Transmission] by Karl-Dirk Kammeyer, 2nd edition, B. G. Teubner, Stuttgart, ISBN 3-519-16142-7 (FIG. 12.1.7 on page 417). A disadvantage in this case is the high realization cost, which arises, on the one hand, from the necessary carrier control circuit with the associated high-rate (higher than the chip rate) multiplication of the receive signal with the frequency- and phase-accurate carrier wave and, on the other, from the costly and complex signal processing with a high-rate complex matched filtering. This high realization cost in addition causes a very high power consumption.
An incoherent receiving unit is known furthermore from the aforementioned textbook (FIG. 12.3.7 on page 447). It has an FM discriminator, an integration unit, and a so-called limiter and requires the processing of high-rate (higher than the chip rate) and sometimes complex-valued signals. This is associated in turn with a high realization cost and a high power consumption. In addition, the efficiency (symbol error rate, etc.) of this receiving unit during demodulation of MSK signals is not adequate.
For the detection of data symbols contained in a receive signal or for the detection of data symbol boundaries, the transmit symbols to be transmitted are typically transmitted within transmission frames, whereby a sequence, e.g., a PN sequence, known receive-side, is appended in the form of a preamble to each transmission frame. Frame detection, during which the symbol boundaries are determined, is performed first in the receiver based on the preamble. A conventional receive signal is shown in FIG. 3. It comprises, for example, transmit symbols 301 and 303 each with L sampling values (coefficients) and a preamble 305, which contains, e.g., the coefficients 010110, already known receive-side, and repeats.
For receive-side frame synchronization, the receive signal is first supplied to cross-correlation filter 401 (KKF), which is shown in FIG. 4 and performs a cross correlation between the receive signal and the preamble word or preamble symbol (preamble). The output signal of cross-correlation filter 401 has periodic peaks, which in each case indicate a correlation maximum. A correlation maximum arises during complete or almost complete overlapping of the preamble in the receive signal and the preamble used receive-side for cross correlation. For this reason, a conclusion can be reached about the particular frame or symbol boundary based on the correlation maxima, which can be detected, for example, by means of a threshold value detector.
Based on the channel properties, such as, for example, multipath propagation or channel noise, the correlation maxima at the output of cross-correlation filter 401, however, are relatively weakly pronounced. In order to express the correlation maxima more strongly, a comb filter 403, which is, for example, an IIR filter with a low-pass characteristic (IIR: Infinite Impulse Response), can be connected downstream of cross-correlation filter 401. An improved correlation signal forms at the output of filter 403; as shown in FIG. 4, it has clearly pronounced correlation maxima and reduced correlation minima in comparison with the correlation signal at the output of cross-correlation filter 401. The time position of the correlation maxima remains unchanged here.
FIG. 5 shows a block diagram of the correlation device of FIG. 4 with a correlation filter 501, which is an FIR filter (FIR: Finite Impulse Response), and an IIR filter, which is connected downstream to the FIR filter 501 and which has an adder 503 and a delay element 505 disposed in a feedback loop and producing a delay by an L clock. Further, amplifying elements 507 and 509 are provided, which also belong to the IIR filter, to amplify the signals applied at the inputs of adder 503. The output signal of the adder provides a sequence on whose basis the frame detection can be performed.
The structure of correlation filter 501 of FIG. 5 is shown in greater detail in FIG. 6. The correlation filter comprises a shift register 601 for delaying the input signal sequence by an L clock, whereby shift register 601 further has a number of outputs to provide the content of the register cells coefficient-wise via the number of register outputs. Multipliers 603 are connected downstream to the register outputs and perform a coefficient-wise multiplication of the register coefficients with already known, receive-side preamble coefficients. The multiplication results are summed up by means of an addition element 605, whereby the summation result is supplied to adder 503 via amplifier 507. As a result, a convolution with a temporally twisted correlation sequence is performed.
A disadvantage of the correlation device shown in FIG. 6 is its high complexity and the associated increased power requirement, because two delay elements, 505 and 601, are needed for its realization.